UCC2975, UCC2976, UCC2977

UCC3975, UCC3976, UCC3977

SLUS499A NOVEMBER 2001 REVISED JANUARY 2002

MULTI TOPOLOGY PIEZOELECTRIC

TRANSFORMER CONTROLLER

www.ti.com

FEATURES

3-V to 13.5-V Operation

Supports Flyback (UCC3975), Half-Bridge
(UCC3976), and Push-Pull (UCC3977)
Topologies

Programmable Voltage Controlled Oscillator

Open Lamp Protection

Low Shutdown Current (15-

A Typical)

Dual MOSFET Drivers

8-Pin TSSOP package

APPLICATIONS

Notebook Computers

Portable Electronics Displays

Portable Instruments

DESCRIPTION

Liquid crystal display (LCD) enclosures and cold
cathode fluorescent lamps (CCFLs) used in
notebook computer and portable electronics
displays are becoming increasingly narrow,
generating the need for a low profile CCFL power
supply. Recent advances in single- and
multi-layered piezoelectric ceramic transformers
(PZT) have enabled the development of a new
generation of efficient, size-reduced backlight
converters. The UCC3975/6/7 family of 8-pin PZT
controllers integrate the necessary circuitry for
operating a PZT-based backlight supply using a
flyback, half-bridge, or push-pull topology. The
choice of power topology depends on application
requirements such as input voltage, lamp voltage,
and PZT gain.

RANGE

OPEN/SD

OSC

COMP

VDD

OUTP

OUTN

GND

CFB

RFB

DFB

CCFL

RCS

COSC

ROSC

LRES

PIEZO

XFMR

RHV

RCNT

VDD

COPEN

ROPEN

SHUTDOWN

UCC3976

DOPEN

UDG01092

Figure 1. UCC3976-Based CCFL Power Supply Using a Resonant Half-Bridge Topology

2002, Texas Instruments Incorporated

PRODUCTION DATA information is current as of publication date.

Products conform to specifications per the terms of Texas Instruments

standard warranty. Production processing does not necessarily include

testing of all parameters.

UCC2975, UCC2976, UCC2977

UCC3975, UCC3976, UCC3977

SLUS499A NOVEMBER 2001 REVISED JANUARY 2002

www.ti.com

description (continued)

A half-bridge PZT converter, using the UCC3976 is shown in Figure 1. External P- and N-channel MOSFETs
are driven out of phase at a fixed 50% duty cycle with anti-cross conduction circuitry provided by the controller.
The half-bridge topology uses only a single magnetic component (LRES) reducing board area. As explained
in the applications section of this datasheet, regulation of lamp current is achieved by controlling the operating
frequency of the system.

The UCC3977 is designed to control a resonant push-pull topology as shown in Figure 2. This controller
alternately drives external N-channel MOSFETs at 50% duty cycle. The push-pull topology requires two external
inductors (L1 and L2), but has the advantage of providing increased voltage across the piezoelectric transformer
primary. In this case a small overlap is provided to the gate drives, assuring an uninterrupted path for inductor
current.

UDG01097

RANGE

OPEN/SD

OSC

COMP

VDD

OUT1

OUT2

GND

CFB

RFB

DFB

CCFL

RCS

COSC

ROSC

RHV

RCNT

VDD

COPEN

ROPEN

DOPEN

SHUTDOWN

UCC3977

PIEZO XFMR

VCNT

Figure 2. UCC3977 Based CCFL Power Supply Using a Resonant Push-Pull Topology

For piezoelectric transformer applications requiring additional gain, a resonant flyback topology can be
implemented using the UCC3975. As shown in Figure 3, a magnetic transformer (T1) provides a stepped up
voltage to the piezoelectric transformer primary. When compared to a traditional high-voltage transformer used
in a CCFL application, T1 is small and low profile because of its reduced turns ratio and voltage rating. In the
resonant flyback application, a single switch is driven at 50% duty cycle producing a half wave rectified sinusoid
at the piezoelectric transformer primary.

UCC2975, UCC2976, UCC2977

UCC3975, UCC3976, UCC3977

SLUS499A NOVEMBER 2001 REVISED JANUARY 2002

www.ti.com

description (continued)

UDG01098

RANGE

OPEN/SD

OSC

COMP

VDD

OUTP

OUTN

GND

CFB

RFB

DFB

CCFL

RCS

COSC

ROSC

RHV

RCNT

VDD

COPEN

ROPEN

SHUTDOWN

UCC3975

VCNT

PIEZO

XFMR

Figure 3. UCC3975-Based CCFL Power Supply Using a Resonant Flyback Topology

absolute maximum ratings over operating free-air temperature (unless otherwise noted)

Supply voltage

VDD

15 V

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Input voltage

OPEN/SD, OSC, COMP, FB, VDD, OUTP

GND0.5 V to V

+0.5 V

. . . . . . . . . . . . . .

Storage temperature range, T

stg

C to 150

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Junction temperature range, T

C to 125

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds

300

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and

functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

§ All voltages are respect to GND.

AVAILABLE OPTIONS

PACKAGED DEVICES TSSOP (PW)

TOPOLOGY

FLYBACK

HALF-BRIDGE

PUSH-PULL

C to 85

UCC2975PW

UCC2976PW

UCC2977PW

C to 70

UCC3975PW

UCC3976PW

UCC3977PW

The PW package is available taped and reeled. Add TR suffix to device type

(e.g. UCC2975TRPW) to order quantities of 2500 devices per reel.

UCC2975, UCC2976, UCC2977

UCC3975, UCC3976, UCC3977

SLUS499A NOVEMBER 2001 REVISED JANUARY 2002

www.ti.com

electrical characteristics V

= 3 V to 13.5 V, T

= 0

C to 70

C for UCC3975/UCC3976/UCC3977,

= 40

C to 85

C for the UCC2975/UCC2976/UCC2977, T

= T

(unless otherwise noted)

input supply

PARAMETER

TEST CONDITIONS

MIN

TYP

MAX

UNITS

VDD supply current

Normal, VDD = 12 V

2.5

VDD supply current

Shutdown

100

VDD UVLO (turn-on) threshold voltage

2.70

2.85

3.00

UVLO hysteresis

100

200

300

output

PARAMETER

TEST CONDITIONS

MIN

TYP

MAX

UNITS

Pchannel driver output voltage, VOUTP

IPIN = 100 mA,

Driving logic low

0.5

0.9

P-channel driver output voltage, (VDD VOUTP)

IPIN = 100 mA,

Driving logic high

0.5

0.9

Low-level Nchannel driver output voltage, VOUTN

IPIN = 100 mA,

Driving logic low

0.5

0.9

High-level Nchannel driver output voltage,
(VDD VOUTN)

IPIN = 100 mA,

Driving logic high

0.5

0.9

Rise time

VDD = 5 V,

CL = 1 nF,

200

Fall time

VDD = 5 V,

CL = 1 nF,

See Note 1

200

Dead (overlap) time

See Note 1

250

oscillator

PARAMETER

TEST CONDITIONS

MIN

TYP

MAX

UNITS

Upper threshold voltage

1.6

1.7

1.8

Lower threshold voltage

0.65

0.70

0.80

Frequency

ROSC = 24 k

, COSC = 470 pF

100

105

kHz

error amplifier

PARAMETER

TEST CONDITIONS

MIN

TYP

MAX

UNITS

Input voltage

1.465

1.500

1.535

Line regulation voltage

3 V

VDD

13.5 V

Input bias current

500

100

Open loop gain

0.5 V

COMP

3.0V,

See Note 1

Low-level output voltage

FB = 2 V,

OPEN/SD = 1 V

ICOMP = 0.23 mA

0.08

0.15

FB = 1 V,

COMP = 2 V

1.5

5.0

Output source current

FB = 1 V,

COMP = 2 V,

OPEN/SD = 3 V

FB = 2 V,

COMP = 2 V

2.5

4.5

Output sink current

FB = 2 V,

COMP = 2 V,

OPEN/SD = 3 V

Unity gain bandwidth frequency

TJ = 25

See Note 1

MHz

NOTE: 1. Ensured by design. Not production tested.

UCC2975, UCC2976, UCC2977

UCC3975, UCC3976, UCC3977

SLUS499A NOVEMBER 2001 REVISED JANUARY 2002

www.ti.com

electrical characteristics V

= 3 V to 13.5 V, T

= 0

C to 70

C for UCC3975/UCC3976/UCC3977,

= 40

C to 85

C for the UCC2975/UCC2976/UCC2977, T

= T

(unless otherwise noted)

mode select

PARAMETER

TEST CONDITIONS

MIN

TYP

MAX

UNITS

Shutdown threshold voltage

2.45

2.50

2.65

Restart threshold voltage

0.3

0.5

0.7

Open lamp detect enable threshold voltage

UCC2975
UCC2976
UCC2977

1.3

1.5

1.6

Open lamp detect enable threshold voltage

UCC3975
UCC3976
UCC3977

1.4

1.5

1.6

MODE pull-down current

200

250

300

No lock threshold voltage

2.4

2.5

2.6

functional block diagram

The UCC397x family of controllers contain an error amplifier whose output is preconditioned at startup, a
precision window comparator used to form the VCO, and dual MOSFET drivers customized for half-bridge or
push-pull operation. The part includes a frequency lock retry circuit, low current shutdown, and open lamp fault
protection.

UDG01053

Figure 4

OPEN/SD

2.5 V

0.5 V

1.5 V

SLEEP

PWRUP

SHUT DOWN

RESET

OPEN LAMP

PWRUP

FAULT

INIT

OSC

COMP

FAULT

COUNTER

RESET

COUNT

FAULT

SLEEP

NO LOCK

INIT

SLEEP

REF

UVLO

2.5 V

0.1 V

INIT

ERROR
AMPLIFIER

1.5 V

1.4 V

1.5 V

1.7 V

0.7 V

1.75 V

DEADTIME/OVERLAP

TIME

GENERA

GND

OUTN/OUT2

OUTP/OUT1

VDD

SLEEP

UCC2975, UCC2976, UCC2977

UCC3975, UCC3976, UCC3977

SLUS499A NOVEMBER 2001 REVISED JANUARY 2002

www.ti.com

Terminal Functions

TERMINAL

NO.

NAME

UCCx975

UCCx976

UCCx977

I/O

DESCRIPTION

COMP

Output of the error amplifier and control voltage used to set the VCO frequency

Inverting input to the error amplifier

GND

Ground reference for the device

OPEN/SD

Open lamp protection and a low power shut down

OSC

Common connection point for components that control the frequency range for the voltage con-
trolled oscillator (VCO)

OUTP/OUT1

Output of an internal CMOS driver used to drive an N-channel MOSFET (for UCC3977), or a
P-channel MOSFET (for UCC3976) left open for UCC3975

OUTN/OUT2

Output of an internal CMOS driver used to drive an N-channel MOSFET.

VDD

Connects to the battery or system voltage

OPEN/SD

OSC

COMP

VDD
OUTP
OUTN
GND

UCC2975, UCC2976
UCC3975, UCC3976

PW PACKAGE

(TOP VIEW)

OPEN/SD

OSC

COMP

VDD
OUT1
OUT2
GND

UCC2977, UCC3977

PW PACKAGE

(TOP VIEW)

UCC2975, UCC2976, UCC2977

UCC3975, UCC3976, UCC3977

SLUS499A NOVEMBER 2001 REVISED JANUARY 2002

www.ti.com

pin assignments

OPEN/SD: This dual-purpose pin provides open lamp protection and a low power shutdown capability for the
part. This pin can also be used to provide burst mode dimming explained in the applications section that follows.

open lamp function

During startup this pin is internally driven low setting the initial condition for the open lamp circuit. An
external peak detection circuit interfaces between this pin and the lamp. If the voltage at the pin exceeds
1.5 V, an open lamp is assumed and the part re-initiates a startup sequence up to 7 times. If the lamp fails
to strike after 7 tries, the device enters an error shutdown mode. An open lamp induced shutdown can be
cleared either by cycling power on the device or by pulling the pin above 2.5 V and then below 0.5 V.

shutdown function

The device is put into shutdown mode (15-

A of typical quiescent current) by forcing the pin above 2.5 V.

When this pin is subsequently brought below 0.5 V, the device comes out of shutdown mode and initiates
a new startup cycle. This pin can be used to delay startup until the system voltage is sufficient to strike
and operate the piezoelectric transformer.

OSC: This pin is the common connection point for components that control the frequency range for the voltage
controlled oscillator (VCO). An external RC circuit connected from this pin to ground sets the center frequency
for the VCO, where a second resistor connected from this pin to the COMP pin sets the allowable frequency
range. A precision window comparator is used to keep the exponentially decaying ramp voltage at this pin
between 0.7 V and 1.7 V. When the voltage decays below 0.7 V, an internal pull-up circuit charges this pin to
1.7 V, the voltage is then allowed to decay to 0.7 V at a rate determined by the external components. Equations
are provided in the applications section to assist in determining the size of the external components to achieve
the desired frequency range.

COMP: This pin is the output of the error amplifier and control voltage used to set the VCO frequency. During
startup internal switches precondition this output to 0 V producing the maximum frequency of operation. The
error amplifier is then allowed to slew its output voltage until the lamp strikes and lamp current is regulated. The
slew rate is set by the external feedback components. If this pin reaches 2.5 V, regulation was not achieved and
startup will be reinitiated up to 7 times.

FB: This is the inverting input to the error amplifier. This input is compared to 1.5 V and is used to control lamp
current.

OUTP/OUT1: This pin is the output of an internal CMOS driver used to drive an N-channel MOSFET in the case
of the UCC3977 or a P-channel MOSFET in the case of the UCC3976. This pin is low slightly less than 50%
duty cycle in the case of the UCC3976 to prevent cross-conduction and is high slightly more than 50% duty cycle
in the case of the UCC3977 to provide overlap. This pin is left open for the UCC3975.

OUTN/OUT2: This pin is the output of an internal CMOS driver used to drive an N-channel MOSFET in the case
of the UCC3975 and UCC3976 or the second N-channel MOSFET in the case of the UCC3977. This pin is high
slightly less than 50% duty cycle in the case of the UCC3975 and UCC3976. The pin is high slightly more than
50% duty cycle in the case of the UCC3977 to provide overlap.

VDD: This pin connects to the battery or system voltage. This pin should be bypassed with a minimum of 0.1-

of capacitance directly at the device , with an additional 5-

F to 10-

F low ESR bulk capacitor (ceramic is

preferred).

GND: Ground reference for the device. This pin should be used as the common ground point for power and
signal level ground traces.

UCC2975, UCC2976, UCC2977

UCC3975, UCC3976, UCC3977

SLUS499A NOVEMBER 2001 REVISED JANUARY 2002

www.ti.com

state diagram

A logic state diagram for the UCC397x family of controllers is shown in Figure 5. During power-up the controller
transitions from UVLO to the momentary startup state. During startup, the COMP pin is preconditioned at
maximum frequency and the OPEN/SD capacitor is discharged before beginning normal operation. In the
normal operating state, the frequency is swept from high to low allowing the lamp to be struck and the current
in the lamp to be regulated.

STARTUP *

OPEN/SD = 0V

COMP = 0V

OSC, OUT active

NORMAL OPERATION

OPEN/SD < 1.5V

COMP < 2.5V

OSC, OUT active

OPEN LAMP *

1.5V < OPEN/SD <2.5V

COMP < 2.5V

OSC, OUT active

NOLOCK *

COMP > 2.5V

OPEN/SD < 1.5V
OSC, OUT active

COUNT=7?

ERROR

SHUTDOWN

OSC active

OUT off state

UVLO

VDD < 3.0V

OSC inactive

OUT off state

COUNT=0

SHUTDOWN

OPEN/SD > 2.5V

COUNT =0

OSC inactive

OUT off state

Low current VDD,

OPEN/SD

VDD>3.0V

OPEN/SD

< 0.5V

YES

Increment COUNT

OUT off state

UCC3975: OUT1 low, OUT2 low

UCC3976: OUTP high, OUTN high

UCC3977: OUT1 low, OUT2 low

* Momentary states

UDG01102

Figure 5. State Diagram

The normal operating state can be exited in one of four ways:

bringing V

< 3 V

a user commanded shutdown (OPEN/SD > 2.5 V)

an open lamp condition (OPEN/SD > 1.5 V), or

if the device fails to achieve regulation before reaching minimum frequency (EAO > 2.5 V).

The latter two conditions cause an internal retry counter to increment before attempting another startup. If the
application does not operate normally after seven retrys, the controller enters an error induced shutdown state
removing power to the load. The error state and counter can be cleared by removing V

to the part or by a

user commanded shutdown.

UCC2975, UCC2976, UCC2977

UCC3975, UCC3976, UCC3977

SLUS499A NOVEMBER 2001 REVISED JANUARY 2002

www.ti.com

APPLICATION INFORMATION

PZT operation

Ceramic piezoelectric transformers were first proposed by C.A. Rosen in 1956. Unlike magnetic transformers
that rely on electromagnetic energy transfer, PZTs transfer electric potential to mechanical force as shown in
Figure 6. The electrical-to-mechanical conversion of energy is referred to as the reverse piezoelectric effect
whereas the mechanical-to-electrical energy conversion is referred to as the direct piezoelectric effect.

ELECTRIC

POTENTIAL

ELECTRIC

POTENTIAL

MECHANICAL

FORCE

MECHANICAL

FORCE

UDG01099

Figure 6. Piezoelectric Effect

Each manufacturer has a unique recipe of materials and structural layering that determine their PZT's operating
characteristics. Common materials used to make PZTs include lead zirconate, lead titanate and lithium niobate.
Single layer PZTs are less costly and easier to manufacture, but have a lower voltage gain (typically 5 to 10 )
and may require a step-up magnetic transformer in order to operate the lamp. Multi-layered PZT designs are
more difficult to manufacture, but have a higher voltage gain (20 to 70) allowing a CCFL to be driven using
conventional off-the-shelf inductors.

UCC2975, UCC2976, UCC2977

UCC3975, UCC3976, UCC3977

SLUS499A NOVEMBER 2001 REVISED JANUARY 2002

www.ti.com

APPLICATION INFORMATION

MECHANICAL

DISPLACEMENT

MECHANICAL

STRESS

VIN

VOUT

SUPPORTS

L = LENGTH

T = THICKNESS

h = HEIGHT

FORCE

PRIMARY

SECONDARY

UDG01076

Figure 7. Typical Longitudinal Mode Piezoelectric Transformer for CCFL Applications.

A typical multi-layer PZT with longitudinal mode geometry is shown in Figure 7, a single layer design would have
similar construction without the layering on the primary. An ac voltage is applied to the V

electrodes causing

mechanical expansion and compression in the thickness direction (see Figure 6). This displacement on the
primary is transferred as a force in the longitudinal direction. Supports at

and

wavelength provide a means

for a standing wave to be generated at a resonant frequency as shown. Mechanical resonance occurs at
multiple standing wave frequencies (n) based on the transformer's length and material velocity (v).

length

Voltage gain is a function of the PZT material coefficient g[

], the number of primary layers, the thickness of

the material and the overall length as follows:

GAIN

length

layers

thickness

]

An electrode at V

OUT

is used to recover the amplified electrical potential at the secondary.

PZT electrical model

In order to predict PZT performance in a system, it is useful to develop an electrical circuit model. The model
shown in Figure 8 is often used to describe the behavior of a PZT near the fundamental resonant frequency.
Many PZT manufacturers will provide component values for the model based on measurements taken at
various frequencies and output loads.

(1)

(2)

UCC2975, UCC2976, UCC2977

UCC3975, UCC3976, UCC3977

SLUS499A NOVEMBER 2001 REVISED JANUARY 2002

www.ti.com

APPLICATION INFORMATION

1 : n

VIN

CINPUT

COUT VOUT

RLOAD

UDG01100

Figure 8. Equivalent Piezoelectric Transformer Circuit Model

The component values depend on the PZT's construction. A large primary capacitance (C

INPUT

) is formed as

a result of the multi-layer construction of the primary electrodes and material dielectric constant. The output
capacitance is much smaller due to the distance between the primary and secondary electrodes. Typical values
of C

INPUT

and C

OUT

for a multilayer PZT may be 0.2

F and 20 pF respectively, where a single layer design

would have lower C

INPUT

since layers =1.

INPUT

length

width

layers

thickness

OUTPUT

thickness

width

length

INPUT

and an external transformer or inductor(s) are used to form a primary-side L-C resonant circuit as

depicted in Figures 1, 2 and 3. These circuits provide sinusoidal waveforms at the primary, allowing the PZT
to operate at higher efficiency. The mechanical resonant frequency (

) of the PZT (which differs from the

natural primary L-C resonant frequency) is proportional to the material elasticity (Y), density (

) and length.

length

The mechanical piezoelectric gain near a single resonant frequency can modeled by a series R, L, and C circuit
as depicted in Figure 8.

+ w

Figure 9 illustrates the gain-vs-output load and frequency characteristics for a 12-layer, 70-kHz PZT with the
following Figure 8 values:

INPUT

= 0.2

OUT

= 30 pF

n = 30

series RLC (2

, 1

H, 6 nF)

As shown in Figure 9, the ceramic transformer provides high Q and gain under light or no-load conditions
producing a high-strike potential. Once the lamp strikes the transformer becomes loaded, causing the
transformer gain to decrease and resonant frequency to shift. The piezoelectric transformer is typically operated
on the right side of resonance to allow the lamp to be struck and operated with a single direction control circuit.
A typical application has separate start (A), strike (B), and operating (C) frequencies (see Figure 9).

(3)

(4)

(5)

(6)

(7)

UCC2975, UCC2976, UCC2977

UCC3975, UCC3976, UCC3977

SLUS499A NOVEMBER 2001 REVISED JANUARY 2002

www.ti.com

APPLICATION INFORMATION

lamp characteristics

A cold cathode fluorescent lamp has non-linear V-I characteristics as shown in Figure 10. The lamp's intensity
(lumens) is roughly proportional to lamp current where lamp voltage remains somewhat constant over the
operating range. Lamp voltage is dependant on the diameter and length of the lamp used in the application. This
results in increased impedance when the lamp is dimmed. The impedance of the lamp will influence the gain
of the piezoelectric transformer (see Figure 9) and thus the operating frequency of the system.

start

Figure 9

150

100

200

300

250

ROUT = 500 k

ROUT = 250 k

ROUT = 100 k

ROUT = 750 k

f Frequency kHz

GAIN

Gain

PIEZOELECTRIC GAIN

FREQUENCY

strike

operate

Figure 10

550

500

650

600

700

800

750

500

1500

1000

2000

3000

2500

Impedance

LAMP

Lamp V

ltage

CCFL CURRENT

LAMP VOLTAGE AND IMPEDANCE

ILAMP Lamp Current mA

LAMP

VOLTAGE

LAMP

IMPEDANCE

variable frequency control system

A simplified block diagram of a PZT based backlight converter is shown in Figure 11. The PZT is driven by a
resonant power stage whose amplitude is proportional to input voltage. The PZT then provides the voltage gain
necessary to drive the lamp. A control loop is formed around the error amplifier that compares average lamp
current to a reference signal (REF) allowing the intensity of the lamp to be regulated. The resulting control
voltage V

drives a VCO that determines the operating frequency of the resonant power stage.

The frequency range of the VCO must include the strike and operating frequencies of the PZT with some
tolerance included for component variation. Minimizing the programmable range improves the control response
of the feedback loop. For example, a frequency range of 67 kHz to 77 kHz might be used for the PZT in Figure 9.
The gain of the PZT must guarantee sufficient lamp voltage at minimum input voltage to ensure that the control
loop will always operate on the right side of resonance.

UCC2975, UCC2976, UCC2977

UCC3975, UCC3976, UCC3977

SLUS499A NOVEMBER 2001 REVISED JANUARY 2002

www.ti.com

APPLICATION INFORMATION

RESONANT

POWER

STAGE

DC INPUT
VOLTAGE

VOLTAGE

CONTROLLED

OSCILLATOR

REF

CCFL

LAMP CURRENT

SENSE

ERROR

AMPLIFIER

PIEZOELECTRIC

TRANSFORMER

UDG01101

Figure 11. Control System for Variable Frequency PZT Backlight Control

programming the frequency range

The frequency range of the UCC397x family is programmed with external components R

OSC

, C

OSC

, and R

ANGE

(see Figures 2 and 3). The programmed range should include the strike and operating frequency required for
lamp operation, plus sufficient tolerance for component variations. An accurate NPO capacitor is recommended
for C

OSC

(between 100 pF and 1000 pF) while 1% resistors are recommended for R

OSC

and R

ANGE

. The VCO

frequency is determined by the charge and decay times between 0.7 V and 1.7 V at the OSC pin. When the
voltage reaches 0.7 V, an internal pull-up circuit charges OSC back to 1.7 V, the charge time (t

CHG

) varies with

the value of C

OSC

but is typically on the order of 500 ns. The decay time (t

DISCH

) is determined by the value

of C

OSC

and the discharge currents generated in R

OSC

and R

ANGE

. The nominal discharge time at OSC is set

by R

OSC

and C

OSC

(see equation [8]), the frequency range is programmed by adjusting the discharge time with

the R

ANGE

resistor and the COMP voltage (see equation [9]):

nominal

DISCH

OSC

ln 1.7

0.7

with lamp

DISCH

COMP

OSC

ANGE

OSC

)

ANGE

1.7 R

OSC

)

ANGE

COMP

OSC

0.7 R

OSC

)

ANGE

COMP

OSC

resulting frequency

Frequency V

COMP

CHG

)

DISCH

COMP

(8)

(9)

(10)

UCC2975, UCC2976, UCC2977

UCC3975, UCC3976, UCC3977

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APPLICATION INFORMATION

Equations 8 and 9 are derived by solving Laplace or differential equations for the RC decay time from 1.7 V to
0.7 V with and without the effect of V

COMP

. The resulting frequency of the system is given in equation 10. This

frequency should be verified in the lab and may need adjustment depending on factors such as extra
capacitance at the OSC pin (oscilloscope measurements can affect frequency) as well as noise. Figure 12
shows the resulting frequency-to-control voltage relationship with the component values listed below the figure
and a t

CHG

time of 500 ns.

Figure 12

f OSC

Oscillator Frequency

kHz

ERROR AMPLIFIER VOLTAGE

OSCILLATOR FREQUENCY

VCOMP Error Amplifier Voltage V

0.5

1.0

1.5

2.0

2.5

ROSC = 15.8 k

COSC = 560 pF

RANGE = 162 k

UCC2975, UCC2976, UCC2977

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SLUS499A NOVEMBER 2001 REVISED JANUARY 2002

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APPLICATION INFORMATION

setting lamp current

The lamp current is controlled by adjusting the frequency of the PZT. System frequency and lamp current control
is accomplished through the error amplifier (EA) and the voltage controlled oscillator (VCO) as shown in
Figure 12. Lamp current is sensed at RCS and is averaged at EA by RFB and CFB.

LAMP

RCS

Equation (11) assumes the error amplifier loop is closed, the relationship between V

and V

CNT

(dimming

control voltage) is given in equation (4).

1.5 V

CNT

)

CNT

)

CNT

The relationship between control voltage and lamp current can be easily programmed for the application. For
example suppose maximum lamp current is 5 mA (V

CNT

= 0 V) and minimum lamp current is 1 mA

CNT

= 3 V). R

is calculated to be 1100

by using equation (12) and setting the lamp current to 3 mA

CNT

= 1.5 V, V

= 1.5 V). R

CNT

is calculated to be 150 k

by selecting R

at 100 k

and solving equation

(12) at maximum lamp current (V

CNT

= 0 V, I

LAMP

= 5 mA). Using these, the resulting control voltage to lamp

current equation becomes:

LAMP

3.75

CNT

742

= 1100

CNT

= 150 k

= 100 k

sizing the feedback capacitor

Feedback design with a PZT requires both modeling and measurement. The uncompensated feedback gain
for the system is primarily affected by the gain slope of the PZT near its resonant operating frequency as shown
in Figure 9. For most designs, the safe unity gain crossover frequency of the feedback loop will be determined
by the amount of gain peaking that occurs at the resonant frequency of the PZT transformer. R

and C

are

selected to have a fairly low crossover frequency to ensure that the system gain does not increase above unity
at the resonant switching frequency. Since the gain slope is dependant on the lamp load and PZT model, it is
recommended that a network analyzer is used to validate sufficient gain and phase margin for the design.

A simple first order (or integral) feedback stage is used to stabilize the feedback response of the system.
Selection of the feedback capacitor (C

) and resistor (R

) is primarily dependant upon the small signal gain

of the system and the desired sweep rate of the VCO. If the frequency is swept too rapidly at startup (with an
undersized C

), the feedback loop will not stabilize after the lamp is struck and the controller will cycle through

the VCO frequency range without locking. A feedback capacitor that is too large has poor transient performance.

A C

value of 0.1

F is usually a good starting point for most designs if R

is 100 k

. With analog dimming,

the C

value must be large enough to be stable at high V

and minimum lamp current (maximum PZT gain

slope and load). The C

value can be decreased with burst dimming since the lamp is operated at full load

where the PZT gain slope is reduced.

(11)

(12)

(13)

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APPLICATION INFORMATION

matching the PZT to the lamp, input voltage and topology

A fundamental challenge in the design of a piezoelectric transformer-based CCFL application is to match the
lamp's power requirements with the transformer. Since the piezoelectric transformer is a mechanical system,
the energy delivered by the transformer is a function of its mass and its vibrational velocity (

energy

mass

The power delivered by the transformer is described in equation (15):

power

energy

frequency

The design challenge becomes how operate the transformer within its gain and power delivering capabilities
while optimizing overall system efficiency. This optimization requires knowledge of both the lamp and
piezoelectric transformer for the particular application. Achieving optimal efficiency with a given lamp and
piezoelectric transformer will require bench measurements and design iterations. There are several factors that
should be taken into consideration when selecting a piezoelectric transformer:

What is the recommended input voltage for the PZT?

What is the input capacitance of the PZT?

What is the gain of the transformer at various load conditions? (see Figure 9)

At what frequency does the PZT give maximum gain?

The recommended input voltage and gain of the piezoelectric transformer influence the power topology
selection. As mentioned earlier, the half-bridge topology gives the least gain where the push-pull topology
doubles the primary voltage. The flyback topology can provide additional gain through the flyback transformer.

The input capacitance and operating frequency of the piezoelectric transformer determines the required value
of the external inductor(s) (or transformer inductance in the case of the flyback). The external inductance value
may need to be further optimized to get the best performance.

(14)

(15)

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half-bridge operation and inductor selection

In the half-bridge topology, the external inductor and piezoelectric capacitance form a low-pass filter between
the common switch node of the external MOSFETs and the piezoelectric primary as shown on the front page.
The L-C filter formed by these components should pass the resonant frequency, required by the piezoelectric
transformer, yet attenuate higher harmonic components. The choice of inductor will require bench
measurements and modeling of the resonant circuit:

An inductor value that is too low (high L-C resonant frequency) will result in non-sinusoidal primary
waveforms since higher order harmonics are allowed though the filter. A low value also allows excess
circulating currents, impacting efficiency.

An inductor value that results in a L-C resonant frequency close to the resonant frequency of the
piezoelectric transformer causes interference, making control of the primary voltage difficult. The
interference occurs since the gain of the L-C tank depends heavily on load in this region of operation.

An inductor value that is too large causes an attenuation of the input voltage, increasing the gain
requirements of the piezoelectric transformer and/or the system.

As an example, suppose a piezoelectric transformer is selected that operates efficiently at 67 kHz (similar to
Figure 9) and has 0.2-

F of primary capacitance. An external inductance value of 15

H gives a L-C filter corner

frequency of 92 kHz. The L-C circuit would provide little attenuation at 67 kHz yet attenuate higher harmonics.

Figure 13

f Frequency kHz

GAIN

Gain

UCC3976 L-C TANK FREQUENCY

LAMP LOADS

150

200

250

300

100

1.5

2.0

0.5

fPZT = 67 kHz

LAMP LOAD

150 k

100 k

50 k

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half-bridge operation and inductor selection (continued)

Waveforms for the UCC3976 half-bridge circuit operated with a 12-Vdc input are shown in Figure 14. P- and
N-Channel MOSFETs are driven out of phase with 50% duty cycle producing a square wave at the drains (see
Figure15: trace 1). Inductor LRES and the PZT primary capacitance form a low-pass filter. The resulting in a
near sinusoid across the PZT primary (trace 2). Due to the high Q of the PZT, lamp voltage (trace 4) is sinusoidal.
Lamp current is sensed by the half-wave rectification circuit at RCS (trace 3). Lamp current is in phase with lamp
current since the load is primarily resistive with some capacitance due to the reflector's proximity to the lamp.
The lamp reflector should be grounded for safety reasons and in order to keep the secondary capacitance
constant allowing the PZT load to be constant.

VIN = 12 Vdc

VLAMP = 600 V

Figure 14. UCC3976 Half-Bridge Waveforms

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push-pull operation and inductor selection

For the push-pull circuit, MOSFETs N1 and N2 are driven out of phase with 50% duty cycle at variable frequency
(see Figure 15: trace 2). Inductors L1 and L2 resonate with the PZT primary capacitance, forming a half
sinusoids at the drain of N1 (trace 1) and S2 (trace 4). The resulting voltage across the PZT primary is a near
sinusoid (trace M1). Due to the high Q of the ceramic transformer, the lamp voltage, which is approximately
600 V in this application, is sinusoidal (trace 3). In order to achieve zero-voltage switching, each drain voltage
must return to zero before the next switching cycle. This dictates that the L-C resonant frequency be greater
than the switching frequency. The maximum inductance to meet these conditions can be found from
equation (16):

VIN = 7 Vdc

VLAMP = 600 V

Figure 15. UCC3977 Push-Pull Waveforms

(16)

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APPLICATION INFORMATION

flyback operation

For single layer PZT applications requiring additional gain, a resonant flyback topology can be implemented as
shown in Figure 16. In the resonant flyback application, a single N-channel switch is driven at 50% duty cycle
producing a half sinusoid at the drain (see Figure 16: trace 1). The magnetic transformer provides a stepped
up voltage to the PZT primary (trace 4). The resulting lamp voltage at the PZT secondary, which is approximately
250 V in this case, is sinusoidal resulting from the high Q of the ceramic transformer (trace 3). When compared
to a high-voltage magnetic CCFL transformer, the magnetic step-up transformer is small and low profile
because of the reduced turns ratio (3.5:1) and voltage rating. To ensure zero-voltage switching, as in the case
of the push-pull converter, equation (16) must be validated. The L-C relationship can be analyzed on either the
primary or secondary side of the magnetic transformer. If viewed from the primary, PZT capacitance is reflected
by the turns ratio.

VIN = 4 Vdc

VLAMP = 250 V

Figure 16. UCC3975 Resonant Flyback Waveforms

UCC2975, UCC2976, UCC2977

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analog dimming PZT performance

High efficiency can be achieved by selecting the best power topology while matching the lamp, input voltage
and PZT characteristics. Figure 17 shows the performance of a 3-W rated multi-layer PZT operating a 600 V
lamp using the push-pull topology at various input voltage and lamp current conditions. Electrical efficiency is
greater than 85% at lower input voltages, decreasing at higher input voltages as the PZT gain is reduced. This
circuit and lamp can operate from 2 Li-Ion cells with voltages between 5 V and 8.2 V. The same PZT and lamp
would require three LiIon cells for the half-bridge topology but would yield similar efficiency.

Dimming by linearly reducing lamp current causes the efficiency to degrade since the PZT is operated at less
than optimal gain (see 1.5 mA curve). Improved efficiency can be achieved by using burst mode dimming. This
dimming method involves running the lamp at full power, but controlling average lamp current by modulating
the on/off duty cycle at a frequency higher than the eye can detect (100 Hz, for example).

Figure 18 shows plots of PZT operating frequency over the same lamp conditions as Figure 17. As expected,
frequency decreases at higher lamp currents as the PZT characteristics shift to a lower operating frequency
when loaded (see Figure 2). Frequency increases linearly with input voltage, since the required V

OUT

gain

to operate the lamp is decreased.

Figure 17

TYPICAL PIEZO TRANSFORMER EFFICIENCY

INPUT VOLTAGE

Efficiency

VIN Input Voltage Vdc

1.5 mA

660 V

4.5 mA

570 V

3.0 mA

610 V

Figure 18

f OSC

Oscillator Frequency

kHz

PIEZO TRANSFORMER FREQUENCY

INPUT VOLTAGE

60.5

60.0

62.5

61.0

63.5

64.5

62.0

61.5

64.0

63.0

65.0

VIN Input Voltage Vdc

4.5 mA

570 V

1.5 mA

570 V

3.0 mA

610 V

UCC2975, UCC2976, UCC2977

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open lamp shutdown/no-lock operation

Due to the high gain characteristics of the piezoelectric transformer, it is important that the operation of the power
stage is suspended if an open lamp occurs. Figure 19 shows the output voltage of a piezoelectric transformer
with no output load and driven with a 5-V

RMS

sinusoid on the primary. The primary frequency is swept through

the resonant frequency of the piezoelectric transformer. As seen in Figure 19 the output voltage approaches
2 kV

RMS

with open lamp disabled, a gain of 400! If the input voltage were increased to 10 V

RMS

, the output would

reach 4 kV

RMS

and possibly crack the PZT transformer.

OPEN LAMP

PROTECTION

DISABLED

Maximum

Frequency

Minimum

Frequency

PZT Resonant

Frequency

Figure 19. Damaging Voltages at Piezoelectric Transformer Secondary

In order to prevent damaging voltages at the piezoelectric transformer secondary, a 1.5-V comparator at the
OPEN/SD pin is used to shutdown the converter if an open lamp is detected. The RMS secondary voltage at
which an open lamp fault is triggered can be calculated using equation (17).

OPEN

1.5

)

DIODE

OPEN

will typically consist of several inexpensive high impedance resistors to minimize current in the divider and

to standoff high voltage. R

can be replaced with a single high voltage capacitor if component count is an

issue, however, impedance of the capacitor over frequency needs to be taken into consideration.

(17)

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open lamp shutdown/no-lock operation (continued)

Figure 20 shows the startup performance of the UCC397x family based system with a broken or open lamp.
The open lamp trip level is typically set 20% to 50% higher than the required strike voltage of the lamp, in this
example the open lamp trip level is set at a low 500 V

RMS

. As seen in Figure 21, the lamp does not strike before

the OPEN/SD pin reaches 1.5 V indicating an open lamp, the controller retries seven times before entering an
error shutdown state (see the state diagram in the pin description section).

VCOMP

VOPEN/SD

VPZT

Figure 20. Start-Up With Open Lamp

A second type of failure mode occurs if the system fails to control lamp current. Assuming a proper feedback
network, this failure can occur if the input voltage is too low to operate the lamp or if one of the components in
the power path is open, shorted, or broken. These failures are detected at the COMP pin. Figure 21 shows the
system response where the input voltage is not sufficient for lamp operation. At startup the frequency sweeps
through the range until COMP reaches 2.5 V, the controller attempts seven retries before entering the error
shutdown state. Notice the slope on the COMP pin (trace 2) changes as the system attempts to operate the lamp
in a high gain region but is ultimately unsuccessful.

UCC2975, UCC2976, UCC2977

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burst dimming with the OPEN/SD pin

A burst dimming technique can be used with CCFLs when a wide dimming range is required, this technique can
also yield better efficiency at light loads. Burst dimming is implemented by running the lamp at full current when
on, where the on/off duty factor is controlled a low frequency to provide dimming. In order to prevent visible
flicker, the burst frequency needs to be set higher than 80 Hz.

The UCC397x family is initially targeted for operating the PZT using analog dimming, however, burst dimming
can be implemented by controlling the OPEN/SD pin directly with a square wave. Figure 22 shows burst
dimming performance using the UCC3976 at 125 Hz and 50% duty cycle. In Figures 2225, trace 1 is the drain
connection of the external P- and N-Channel MOSFETs, trace 2 is the OPEN/SD pin which is externally driven
with a 5-V square wave, trace 3 is the COMP pin which is used to lock the operating frequency and trace 4 is
the lamp voltage. These scope graphics were captured with a digital oscilloscope, so aliasing is present in
Figures 2224. Referring back to Figure 22, when OPEN/SD is driven to 5 V the part is in shutdown and the
controller is disabled. When OPEN/SD is forced to 0 V by the external source the controller goes through its
startup sequence with COMP starting at 0 V allowing the PZT to strike the lamp and lock on the frequency
required to regulate the lamp at full current. The size of the feedback capacitor determines the slew rate at which
COMP can lock the system frequency which effects the achievable duty cycle of burst dimming. Fortunately,
the feedback cap with burst dimming can be smaller than with analog dimming since the system small signal
gain is lower at full lamp load. Figures 23 and 24 show burst dimming at approximately 10% and 90% duty cycle
respectively. Figure 25 shows a close up of the startup sequence when OPEN/SD is pulled low. COMP is
preconditioned to 0 V before switching begins and then allowed to ramp up. PZT secondary voltage ramps as
the frequency decreases until the lamp strikes and operates. Strike voltage for the lamp is barely detectable
since the lamp is warm and operating from the previous burst cycles.

VCOMP

VOPEN/SD

VLAMP

VOPEN/SD

Figure 22. UCC3976 Burst Dimming at 50% Duty Cycle

VIN = 12 Vdc

VLAMP = 600 V

fOSC = 125 Hz

50% Duty Cycle
ILAMP = 5 mA

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